The transmission of two complementary signals over identical, matched traces is a special case of two-wire signaling. It is called differential signaling (Figure 6.5). A differential signaling system delivers equal but opposite AC voltages and currents on the two wires. Under these conditions, and assuming the layout is symmetrical so that both wires have equal coupling to the reference system, any AC currents induced in the reference system by one wire are counteracted by equal and opposite signals induced by the complementary wire.

Figure 6.5. Differential transmission: As long as parasitic impedances z 1 and z 2 remain well balanced, no net current flows into the reference system.

This effect is illustrated in Figure 6.5, where parasitic impedances z 1 and z 2 represent the impedance from one wire to the surrounding reference system (chassis or other bits of metal) and from the other wire to the reference system respectively. In a well-balanced differential system these two impedances are equal. As long as the AC voltages on the two wires are complementary, the stray currents through z 1 and z 2 will cancel, resulting in no net flow of stray returning signal current in the reference system .

The performance of such a system does not hinge on the particular value of impedance z A (since no current flows through it). Conversely, the system enjoys a measure of immunity from other circuits that do induce currents in the reference structure, because the differential receiver does need the voltages on the reference system to be the same everywhere. A differential receiver requires only that the disparity between reference voltages at either end of a link not cause the received signal to exceed the common-mode operating range of the receiver.

If the impedances z 1 and z 2 are not well balanced, or if the transmitted voltages are not precisely complementary, some amount of current will flow in the reference system. This current is called common-mode current . Assuming that the transmitted voltages are equal, I'd like to point out two physical means for reducing the magnitude of common-mode current: the weak-coupling approach and the precise-balance approach.

High-performance twisted-pair data cabling (Figure 6.6) uses the weak-coupling approach, whereby the parasitic coupling through z 1 and z 2 to Earth is generally weakened. The weakening is accomplished by thickening the plastic jacket on the cable to keep other wires and objects outside the jacket relatively far away from the signal conductors. At the same time, the cable holds the wires of each pair closely together in a 100-ohm differential configuration. This geometry increases the magnitude of z 1 and z 2 relative to the impedance of the load. As a result, regardless of where the cable is laid, the fraction of current that can possibly be conducted through z 1 or z 2 cannot be very great. Further reductions in the common-mode current are obtained by tightly twisting the wires so that any nearby objects are approached an equal number of times by each wire. The tight twists produce a better balance between z 1 and z 2 . The combination of a weakened coupling magnitude and good balance is used in category 5, 5e, 6, and 7 unshielded twisted-pair cabling to attain spectacular common-mode rejection .

Figure 6.6. Construction of high-performance 4-pair unshielded twisted-pair data cable. The zip cord shown in this figure is a tough string that technicians pull to rip open long sections of the outer insulation.

Lest you doubt that impedances z 1 or z 2 could carry enough signal current to represent a significant radiation problem, let me suggest the following experiment. Beginning with a working LAN adapter, strip back a section of the outer jacket of the twisted-pair cable. Untwist the wires of one of the active pairs, and tape one wire of that pair against a solid metal part of the product chassis for a distance of about two inches. Run in this configuration, the product will not pass its FCC or EN mandated radiation tests. The amount of excess common-mode current flowing through the parasitic capacitance of the one wire taped against the chassis causes enough radiation to exceed FCC or EN mandated limits. Unbalanced stray current to ground makes a big difference in twisted-pair transmision systems and must be strictly limited.

Differential traces on a pcb must take a different approach to the control of common-mode currents. In a pcb with solid plane layers (Figure 6.7) you cannot remove the traces far enough from the solid planes to significantly reduce the magnitude of z 1 or z 2 . Common-mode rejection in solid-plane pcb applications is achieved not by weakening the coupling, but by precisely balancing it. This is done by using traces of equal height, width, thickness , and length.

Figure 6.7. Cross section of pcb layout showing edge- coupled microstrip differential pair. Both lines are symmetrically loaded with respect to the underlying solid reference plane.

In regards to a differential pcb application, it is not particularly important that the traces be coupled tightly to each other. In fact, you couldn't achieve very tight coupling even if you tried. The coupling ratio for typical differential pair on a pcb lies in the 20% to 50% range.

In contrast, for a well-balanced differential twisted pair such as used in the LAN industry, the coupling is excellent . In such a cable if you transmit a fast signal v ( t ) down one wire of the pair, [47] at the far end you will receive two complementary signals, v ( t )/2, and “ v ( t )/2. This effect illustrates the good common-mode rejection of UTP.

[47] At the near end of the cable, transmit voltage v ( t ) between wire A and earth ground, while holding wire B at earth potential. You may do this with a single-ended pulse generator. At the far end of the cable, resistively terminate wires A and B to a common point, and ground that point to your scope. Look at the signals at the far end of wires A and B .

Good twisted-pair cables couple the wires of each pair to each other much better than to the surrounding earth. The crosstalk between the two wires is almost 100%. The pairs affect each other very strongly. The only successful mode of propagation is the differential mode.

In a solid-plane pcb environment, even when the two traces of a differential pair are pressed close together, the coupling between traces still rarely approaches the level of coupling to the surrounding planes. Try it. Working at the near end of a pair of differential traces, impress a signal on trace A , with trace B grounded. Look at the signals at the far end. You'll see a fairly large amount of crosstalk at the far end, but nothing approaching 100%. Differential traces on a pcb won't do that.

Fortunately, differential traces need not be tightly coupled in order to accrue all the main benefits of a differential architecture. Any reasonable spacing will do, provided that the complementary outputs of the transmitter are well balanced and the traces have symmetrical impedances to the nearby power and ground planes. That's all it takes to get immunity to power and ground noise, immunity to ground bounce, and a marked reduction in emissions.

Occasionally you will encounter engineers who insist on placing their differential traces very close together. This arrangement does save space, and it marginally improves the crosstalk performance vis- -vis other nearby signals. What it does not do is magically bullet-proof the system against crosstalk (see Section 6.11.6 "Reducing Local Crosstalk").

With the use of two differential signals, we gain one last important benefit: a reduction in radiated emissions (see Section 6.11.3, "Reducing EMI with Differential Signaling").

POINTS TO REMEMBER

- Differential signaling delivers equal but opposite AC voltages and currents on two wires.
- Assuming the layout is symmetrical, any AC currents induced in the reference system by one wire are counteracted by equal and opposite signals induced by the complementary wire.
- Differential pcb traces need not be tightly coupled to be effective.
- Differential signaling markedly reduces radiated emissions.

Fundamentals

- Impedance of Linear, Time-Invariant, Lumped-Element Circuits
- Power Ratios
- Rules of Scaling
- The Concept of Resonance
- Extra for Experts: Maximal Linear System Response to a Digital Input

Transmission Line Parameters

- Transmission Line Parameters
- Telegraphers Equations
- Derivation of Telegraphers Equations
- Ideal Transmission Line
- DC Resistance
- DC Conductance
- Skin Effect
- Skin-Effect Inductance
- Modeling Internal Impedance
- Concentric-Ring Skin-Effect Model
- Proximity Effect
- Surface Roughness
- Dielectric Effects
- Impedance in Series with the Return Path
- Slow-Wave Mode On-Chip

Performance Regions

- Performance Regions
- Signal Propagation Model
- Hierarchy of Regions
- Necessary Mathematics: Input Impedance and Transfer Function
- Lumped-Element Region
- RC Region
- LC Region (Constant-Loss Region)
- Skin-Effect Region
- Dielectric Loss Region
- Waveguide Dispersion Region
- Summary of Breakpoints Between Regions
- Equivalence Principle for Transmission Media
- Scaling Copper Transmission Media
- Scaling Multimode Fiber-Optic Cables
- Linear Equalization: Long Backplane Trace Example
- Adaptive Equalization: Accelerant Networks Transceiver

Frequency-Domain Modeling

- Frequency-Domain Modeling
- Going Nonlinear
- Approximations to the Fourier Transform
- Discrete Time Mapping
- Other Limitations of the FFT
- Normalizing the Output of an FFT Routine
- Useful Fourier Transform-Pairs
- Effect of Inadequate Sampling Rate
- Implementation of Frequency-Domain Simulation
- Embellishments
- Checking the Output of Your FFT Routine

Pcb (printed-circuit board) Traces

- Pcb (printed-circuit board) Traces
- Pcb Signal Propagation
- Limits to Attainable Distance
- Pcb Noise and Interference
- Pcb Connectors
- Modeling Vias
- The Future of On-Chip Interconnections

Differential Signaling

- Differential Signaling
- Single-Ended Circuits
- Two-Wire Circuits
- Differential Signaling
- Differential and Common-Mode Voltages and Currents
- Differential and Common-Mode Velocity
- Common-Mode Balance
- Common-Mode Range
- Differential to Common-Mode Conversion
- Differential Impedance
- Pcb Configurations
- Pcb Applications
- Intercabinet Applications
- LVDS Signaling

Generic Building-Cabling Standards

- Generic Building-Cabling Standards
- Generic Cabling Architecture
- SNR Budgeting
- Glossary of Cabling Terms
- Preferred Cable Combinations
- FAQ: Building-Cabling Practices
- Crossover Wiring
- Plenum-Rated Cables
- Laying Cables in an Uncooled Attic Space
- FAQ: Older Cable Types

100-Ohm Balanced Twisted-Pair Cabling

- 100-Ohm Balanced Twisted-Pair Cabling
- UTP Signal Propagation
- UTP Transmission Example: 10BASE-T
- UTP Noise and Interference
- UTP Connectors
- Issues with Screening
- Category-3 UTP at Elevated Temperature

150-Ohm STP-A Cabling

- 150-Ohm STP-A Cabling
- 150- W STP-A Signal Propagation
- 150- W STP-A Noise and Interference
- 150- W STP-A: Skew
- 150- W STP-A: Radiation and Safety
- 150- W STP-A: Comparison with UTP
- 150- W STP-A Connectors

Coaxial Cabling

- Coaxial Cabling
- Coaxial Signal Propagation
- Coaxial Cable Noise and Interference
- Coaxial Cable Connectors

Fiber-Optic Cabling

- Fiber-Optic Cabling
- Making Glass Fiber
- Finished Core Specifications
- Cabling the Fiber
- Wavelengths of Operation
- Multimode Glass Fiber-Optic Cabling
- Single-Mode Fiber-Optic Cabling

Clock Distribution

- Clock Distribution
- Extra Fries, Please
- Arithmetic of Clock Skew
- Clock Repeaters
- Stripline vs. Microstrip Delay
- Importance of Terminating Clock Lines
- Effect of Clock Receiver Thresholds
- Effect of Split Termination
- Intentional Delay Adjustments
- Driving Multiple Loads with Source Termination
- Daisy-Chain Clock Distribution
- The Jitters
- Power Supply Filtering for Clock Sources, Repeaters, and PLL Circuits
- Intentional Clock Modulation
- Reduced-Voltage Signaling
- Controlling Crosstalk on Clock Lines
- Reducing Emissions

Time-Domain Simulation Tools and Methods

- Ringing in a New Era
- Signal Integrity Simulation Process
- The Underlying Simulation Engine
- IBIS (I/O Buffer Information Specification)
- IBIS: History and Future Direction
- IBIS: Issues with Interpolation
- IBIS: Issues with SSO Noise
- Nature of EMC Work
- Power and Ground Resonance

Points to Remember

Appendix A. Building a Signal Integrity Department

Appendix B. Calculation of Loss Slope

Appendix C. Two-Port Analysis

- Appendix C. Two-Port Analysis
- Simple Cases Involving Transmission Lines
- Fully Configured Transmission Line
- Complicated Configurations

Appendix D. Accuracy of Pi Model

Appendix E. erf( )

Notes

High-Speed Signal Propagation[c] Advanced Black Magic

ISBN: 013084408X

EAN: N/A

EAN: N/A

Year: 2005

Pages: 163

Pages: 163

- Article 225 Outside Branch Circuits and Feeders
- Article 353 High Density Polyethylene Conduit Type HDPE Conduit
- Article 356 Liquidtight Flexible Nonmetallic Conduit Type LFNC
- Article 502 Class II Locations
- Example No. D10 Feeder Ampacity Determination for Adjustable-Speed Drive Control [See 215.2, 430.24, 620.13, 620.14, 620.61, Tables 430.22(E), and 620.14]

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